Mobile, in particular wireless communication is broadly used in several fields of applications, e.g. in the home, public or office area, as well as for any kind of communication, e.g. speech, data, and/or multi-media communication. Basically, these applications are confronted with two major problems.
First, the available bandwidth for transmitting information is limited due to the general shortage of available spectrum. Accordingly, it is well known to modulate both the amplitude and the phase of the carrier to reduce required bandwidth. For instance, systems, in which the amplitude and the phase are modulated, i.e. which process wide-band complex envelope signals, are EDGE, UMTS (WCDMA), HSxPA, WiMAX (OFDM) and 3G-LTE (OFDM). However, amplifying amplitude modulated carriers without distortion in the transmitter output stage imposes significant linearity constraints on the output stage amplifier.
Second, power efficiency of mobile transmitters is important, since mobile terminals in wireless communication are typical portable and thus, usually battery powered. In such mobile terminals, the output stage of the transmitter unit is the largest power consumer of the whole device. Consequently, any improvement in this stage with respect to power efficiency is important. Known efficient power amplifiers topologies or circuit arrangements are known class-C and class-E radio frequency (RF) amplifiers in which the output amplification devices, e.g. the transistors, conducts current only at the time when the collector-emitter voltage is at its lowest value. Unfortunately, class-C and class-E amplifiers are very nonlinear and introduce substantial distortion of the amplitude modulation. Accordingly, class-C and class-E amplifiers are used mainly in transmitters transmitting signals, which are only modulated in frequency, e.g. where the amplitude, i.e. the “envelope” of the RF carrier is constant. Hence, distortion of the amplitude has no negative effects.
Linear Amplification using Nonlinear Components (LINC) is a well-known concept for high efficient linear power amplification of RF signals. Detailed information may be gathered from S. C. Crips, “Advanced Techniques in RF Power Amplifiers Design”, Artech House 2002, or from D. C. Cox in “Linear Amplification with Nonlinear Components”, IEEE Transactions on Communications, December 1974, pp. 1942-1945.
The LINC concept, also known as out-phasing, is illustrated by means of the simplified out-phasing power amplifier 100 in FIG. 1a. Accordingly, an amplitude and phase modulated RF signal Sin(t) is split in two RF signals S1(t) and S2(t) each being phase modulated and having constant amplitude:
                                          S            in                    ⁡                      (            t            )                          =                              A            ⁡                          (              t              )                                ⁢                      sin            ⁡                          (                                                ω                  ⁢                                                                          ⁢                  t                                +                                  φ                  ⁡                                      (                    t                    )                                                              )                                                          (        1        )                                                      S            1                    ⁡                      (            t            )                          =                              1            2                    ⁢                      sin            ⁡                          (                                                ω                  ⁢                                                                          ⁢                  t                                +                                  φ                  ⁡                                      (                    t                    )                                                  +                                  ψ                  ⁡                                      (                    t                    )                                                              )                                                          (        2        )                                                      S            2                    ⁡                      (            t            )                          =                              1            2                    ⁢                      sin            ⁡                          (                                                ω                  ⁢                                                                          ⁢                  t                                +                                  φ                  ⁡                                      (                    t                    )                                                  -                                  ψ                  ⁡                                      (                    t                    )                                                              )                                                          (        3        )                                          ψ          ⁡                      (            t            )                          =                  arccos          ⁡                      (                          A              ⁡                              (                t                )                                      )                                              (        4        )            
Then, the signals S1(t) and S2(t), having constant amplitudes, can be separately amplified by means of efficient nonlinear saturated power amplifiers (PA) RF PA1 and RF PA2 in amplification branches 110, 120. After the amplification, the output RF signal is reconstructed by means of a signal component combiner. The output signal of the combiner equals the sum (or difference) of the two input signals S1(t) and S2(t):Sout=S′1(t)+S′2(t)=G cos(Ψ(t))sin(ωt+φ(t))=GA(t)sin(ωt+φ(t))  (5),where G represents the gain of the amplification stages, i.e. the power amplifiers RF PA1 and RF PA2.
Ideally voltage sources are to be combined so that the average current in the amplification devices can vary as function of the out-phasing angle Ψ(t). If ideal class-A, class-B or class-C operation is used, the devices act as current sources and the DC current does not vary with the out-phasing angle, meaning that the efficiency will drop linearly with output power, i.e. class-A like. However, in overdriven or saturated class-A, class-B or class-C operation modes, the amplification devices act more as voltage sources. That is, approximately independent of input drive and output current and the DC current is able to vary with the out-phasing angle. Ideally the efficiency will drop according to the square root of output power, i.e. class-B like. So effectively there is no gain in power efficiency compared to a linear class-B PA design.
This afore-mentioned behavior can be explained by the so-called load-pulling effect that one amplifier has on the other. FIG. 1b illustrates, on the right hand side, a simplified equivalent circuit for a differential output signal combiner connected to a saturated power amplifier that is approximated as a set of two voltage generators. The equivalent circuit shows that the reconstruction of the amplitude modulation is accomplished by modulation of the effective load impedance of the amplifiers as function of the out-phasing angle Ψ(t). Accordingly, the load impedance is complex, except for the situation of minimum and maximum output power at Ψ=0° and Ψ=90°. The reactive part of the impedance causes an increase in circulating RF currents, which reduce the efficiency of the combiner and cause the class-B like power efficiency.
It is known that the efficiency may be improved by means of the so-called Chireix output combining technique, which has been described in H. Chireix, “High power out phasing modulation”, Proceedings of the Institute of Radio Engineers (Proc. IRE), vol. 23, no. 11, pp. 1370-1392, November 1935. Basically, Chireix's approach compensates for the reactive part of the effective load impedance by means of fixed compensating reactances, which are connected in parallel to the load as illustrated in FIG. 1c. However practically is it quite difficult to successfully implement the Chireix concept, since in order to make it successful, the input power has to be modulated according to the amplitude modulation (AM) to keep the PA in saturation or the compensation reactances have to be changed dynamically.
U.S. Pat. No. 5,345,189 discloses a circuit for combining first and second signals having the same frequency for use in a high efficiency RF amplification stage wherein the circuit comprises basically only reactive circuit elements and thus, appears to be a resistive load. The first and second signals have a relative phase shift and the circuit generates an output signal proportional to the sum or difference of signals. In one embodiment, the circuit comprises a transformer and two LC circuits in which the capacitance is varied in response to relative phase shift.
Another way to improve the efficiency of a PA is to use true switching-mode amplifiers, i.e. class-D, class-E, class-DE, or class-F operation. From switching-mode amplifiers, the voltage-mode class-D (VMCD) PA comes close to an ideal voltage source and is believed to be suitable for the out-phasing concept. It is noted that VMCD is intended to also mean a single-ended or push-pull class-F type PA, since a class-F PA has similar voltage and current waveforms at the PA output. Several examples have been reported in literature according to the general schematic shown in FIG. 1d from X. Zhang, L. E. Larson and P. M. Asbeck, “Design of Linear Out-phasing Power Amplifiers”, pp. 145-159, Artech House, 2003. FIG. 1d depicts a CMOS implementation of a class-D out-phasing amplifier with transmission line coupler.
A problem associated with this approach is that when using a regular class-D design in combination with a non-coherent Chireix combiner, the problem of the crowbar current, i.e. DC-switching current, and the switching losses due the output capacitance in a class-D amplifier remain unsolved.
M. Tarsia, J. Khoury and V. Boccuzzi, in “A Low Stress 20 dBm Power Amplifier for LINC Transmission with 50% Peak PAE in 0.25 um CMOS,” Proc. 2000 IEEE ESSCIR, pp. 61-64, September 2000 show a class-D PA with adequate dead-time, i.e. a class-DE PA, to eliminate the crow-bar current and overcome the switching losses of the output capacitance. FIG. 1e shows the PA configuration and FIG. 1f shows how two of these PAs are to be used in an out-phasing configuration using a combiner circuit without Chireix components.
A problem with this approach is that when the class-DE PA is optimized as a separate component for optimum zero-voltage switching as proposed by D. C. Hamill, in “Impedance Plane Analysis of Class-DE Amplifier,” IEE Electronics Letters, vol. 30, no. 23, pp. 1905-1906, November 1994, the design parameters are not necessarily optimal for operating this PA in an out-phasing configuration. It was found with simulation that the phase of the output signal after the combiner becomes non-linear with respect to the out-phasing angle, which does not occur when using a traditional class-D design with a duty-cycle of 50%.